Multi-standard multi-rate filter

ABSTRACT

A method is provided for decimating a digital signal by a factor of M and matching it to a desired channel bandwidth. The method applies the digital signal input samples to a (M−1) stage tapped delay line, downsamples the input samples and the output samples of each tapped delay line stage by a factor of M, and applies each of the M downsampled sample value streams to M allpass IIR filters, respectively. The M allpass IIR filtered sample streams are then summed and scaled by a factor of 1/M. The result can then be filtered by a digital channel filter.

CROSS-REFERENCE TO RELATED APPLICATIONS

The present patent application is a continuation of U.S. patentapplication Ser. No. 11/611,542, filed on Dec. 15, 2006 now U.S. Pat.No. 8,176,107, which application is related to and claims priority of(a) U.S. Provisional Patent Application, entitled “Multi-StandardMulti-Rate Filter,” Ser. No. 60/751,437, filed 16 Dec. 2005; and (b)U.S. Provisional Patent Application, entitled “Differential EvolutionDesign Of Polyphase IIR Decimation Filters,” Ser. No. 60/752,619 andfiled on 20 Dec. 2005. Each of the foregoing applications is herebyincorporated by reference in its entirety into the present application.

BACKGROUND OF THE INVENTION

1. Field of the Invention

The present invention relates to a high-performance; small die size, andlow-power dissipation decimation filter.

2. Discussion of the Related Art

Conventional analog filter solutions require switching between differentfilters or different filter components when processing information codedaccording to different standards or different channel bandwidths. Foranalog filters to achieve the required narrow transition bandwidths andhigh stopband attenuations, precise component tolerances are required.Precise component tolerances are difficult to achieve on-chip,necessitating the use of off-chip components, thereby resulting inincreased system cost. Additionally, temperature compensation and agingare also often required.

Conventional digital filter approaches use finite impulse response (FIR)filters or infinite impulse response (IIR) filters. Conventional FIRfilters require large numbers of coefficients to meet the transitionband and stopband attenuation requirements. Further, multiple sets ofthese coefficients are required to support the various coding standardsand channel bandwidths. As a result, a large on-chip memory is required.Conventional IIR filters also require many sections to meet suchrequirements and are sensitive to both coefficient and signalquantization.

For a detailed description of the theory and design of FIR digitalfilters, see Alan Oppenheimer and Ronald Schafer, Digital SignalProcessing (Prentice-Hall 1975), especially chapters 5 and 6. Furtherinformation regarding conventional filter design may also be found in:

-   -   a Lutovac, M. D. and Milic, L. D., “Design of High-Speed IIR        Filters Based on Elliptic Minimal Q-Factors Prototype” (“Lutovac        and Milic”), Conf. ETRAN 2002, Banja Vrucica, pp. 103-106        (2002).    -   b Lutovac, M. D., Tosic, D. V., and Evans, B. L., “Filter Design        for Signal Processing—Using MATLAB and Mathematica”, Prentice        Hall (2001).    -   c Harris, F. J., “Multirate Signal Processing—For Communication        Systems”, Prentice Hall (2004).    -   d Krukowski, A. and Kale, I., “DSP System Design—Complexity        Reduced IIR Filter Implementation for Practical Applications”        (“Krukowski and Kale”), Kluwer Academic Publishers (2003).    -   e Storn, Rainer, “Designing Nonstandard Filters with        Differential Evolution” (“Storn”), IEEE SIGNAL PROCESSING        MAGAZINE, January 2005.

SUMMARY

According to one embodiment of the present invention, a method isprovided for decimating a digital signal by a factor of M and matchingit to a desired channel bandwidth. The method applies the digital signalinput samples to a (M−1) stage tapped delay line, downsamples the inputsamples and the output samples of each tapped delay line stage by afactor of M, and applies each of the M downsampled sample value streamsto M allpass HR filters, respectively. The M allpass IIR filtered samplestreams are then summed and scaled by a factor of 1/M. The result canthen be filtered by a digital channel filter.

The present invention is better understood upon consideration of thedetailed description below and the accompanying drawings.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 shows a functional block diagram of a zero-IF/very low-IF(ZIF/VLIF) radio receiver front-end 100.

FIG. 2 is a graph illustrating aspects of the filter requirements inrelation to the components of the ZIF/VL1F radio front-end.

FIG. 3 is a functional block diagram of IF sampling radio receiverfront-end 300, in accordance with an alternate embodiment of the presentinvention.

FIG. 4 shows an example of an M-path polyphase IIR decimate-by-M filterstructure 400.

FIG. 5 shows half-band, 2-path polyphase IIR decimator structure 500configured in accordance with one embodiment of the present invention.

FIG. 6 shows third-band, 3-path polyphase IIR decimator structure 600,configured in accordance with one embodiment of the present invention.

FIG. 7 shows generalized 2-path polyphase IIR filter structure 700,configured in accordance with one embodiment of the present invention.

FIG. 8 shows respective 1-coefficient (“real”) section and a2-coefficient (“complex”) section IIR allpass filter structures 800 and850, respectively, configured in accordance with embodiments of thepresent invention.

FIG. 9 shows 3-coefficient filter 900 which is formed by cascading 3real sections.

FIG. 10 illustrates quad-ratio (2, 3, 4, and 6) decimator multi-standardfilter 1000, configured in accordance with one embodiment of the presentinvention.

FIG. 11 illustrates quad-ratio (2, 3, 4, and 6) multi-standard decimatorstructure 1100, configured in accordance with yet a further embodimentof the present invention.

FIG. 12 shows an example of the allpass filter sections of H₁ filter1101 of FIG. 11, configured in accordance with the present invention.

FIG. 13 shows an example of the allpass filter sections 1106, 1107 and1108 of H₂ filter 1102, configured in accordance with the presentinvention.

FIG. 14 shows an example of the allpass filter sections 1109, 1110 and1111 of H₃ filter 1102, configured in accordance with the presentinvention.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS

In exemplary embodiments, the present invention provides amulti-standard, multi-ratio decimator with improved performance at alower cost (e.g., smaller die size) and a lower DC power dissipationthan was achieved in previously available decimators. In one suchembodiment, a quad-ratio (e.g., 2, 3, 4, and 6), multi-standard (e.g.,DVB-T/H, ISDB-T and T-DAB) decimator that provides excellent performancewith a minimal number of logic gates and low DC power dissipation isprovided.

The present multi-standard, multi-rate filter trades-off powerdissipation with performance. Decreasing the sampling rate reduces powerdissipation at the expense of increased aliasing distortion. If theadjacent channel interference (ACI) is small, this added distortion maybe acceptable. For example, the sampling rate can be adjusted to asmaller multiple of the orthogonal frequency division multiplexing(OFDM) fast Fourier transform (FFT) sampling rate when the ACI is low,and increased to a higher multiple when the ACI is high. In this waypower dissipation is reduced, when possible, without performancedegradation.

Referring now to FIG. 1, a functional block diagram of zero-IF/verylow-IF (ZIF/VLIF) radio receiver front-end 100 is shown. Multi-standard,multi-rate filters which are configured in accordance with embodimentsof the present invention are provided as the final blocks 101 and 102 ofthe in-phase (I) and quadrature-phase (Q) paths, respectively. Themulti-standard, multirate filters 101 and 102 each have transferfunction H(z) and decimation ratio M. As shown in FIG. 1, signal F_(RF)denotes the input RF signal applied to mixers 103 and 104 for the I andQ paths, respectively. Mixers 103 and 104 also receive as inputs theoutput signal of voltage controlled oscillator (VCO) 105 of frequencyF_(VCO). VCO 105's signal received at mixer 104 in the Q path isphase-shifted from that received at mixer 103 in the I path by π/2. Theoutput signals of mixers 103 and 104 are applied as input signals toanalog low pass filters (LPF) 107 and 108 each of bandwidth B_(LP).Thus, the input RF signal is downconverted to zero 1E, or very-low IF.

The output signals of LPFs 107 and 108 are capacitivly coupled to inputsignals of analog-to-digital (A/D) converters 109 and 110 to remove DCoffsets. The A/D converters sample and quantize their input signals at asampling rate of F_(S). The resulting sampled and quantized signals aredigitally filtered and decimated by multi-standard, multi-rate filters101 and 102 to provide the respective I(k) and Q(k) output signals.

FIG. 2 is a graph illustrating various aspects of the filterrequirements in relation to other components of the ZIF/VLIF radio. Asshown in FIG. 2, frequency F_(PB) denotes the passband bandwidth,frequency F_(SB) is the frequency at the high frequency edge of thestop-band, and frequency F_(S) is the A/D sampling rate, amplitudeA_(PB) is the allowable passband ripple, and amplitude A_(SB) is therequired stopband attenuation.

FIG. 3 is a functional block diagram of IF sampling radio receiverfront-end 300, in accordance with an alternate embodiment of the presentinvention. As shown in FIG. 3, multi-standard multi-rate filters 301 and302 are the final blocks of the in-phase (I) and quadrature-phase (Q)paths, each having transfer function H(z) and decimation ratio M. Inthis embodiment, an input IF signal of frequency F_(IF) (e.g., 36 MHz,36.125 MHz, and 36.17 MHz) is applied to an IF bandpass filter (BPF) 303of bandwidth B_(BP). The resulting bandpass signal is capacitivlycoupled to A/D converter (of sample rate F_(s)) 304 to remove DCoffsets.

A/D converter 304 samples and quantizes the bandpass signal of bandpassfilter 303, and provides as input samples to digital mixers 305 and 306of the I and Q paths. In this case, digital mixers 305 and 306 each alsoreceive as an input digital signal an output digital signal ofnumerically controlled oscillator (NCO) 307, which receives a digitalword F_(NCO), representing the NCO frequency. As in IF sampling receiver100, NCO 307's output signal to the Q path mixer 306 is phase-shifted byπ/2 from that of the input signal to mixer 305 of the I path. Thesampled and quantized IF samples are thus multiplied by the sine andcosine of NCO 307's output phase, respectively. The resulting samplesare digitally filtered, and decimated by multi-standard, multi-ratefilters 301 and 302 to provide the baseband samples.

Digital M-path polyphase infinite impulse response (IIR) filters aresuitable for use in decimators with decimation ratio M, for M greaterthan 1. The M-path polyphase IIR filters are based on an M-tap finiteimpute response (FIR) filter in which the coefficients are replaced byallpass filters. Polyphase IIR filters provide high stopband attenuationand low passband ripple with a relatively small number of coefficients.

FIG. 4 shows an example of an M-path polyphase IIR decimate-by-M filterstructure 400. As shown in FIG. 4, input sample x(n) is applied to anM−1 stage, tapped delay line. The input sample and the output signals ofthe M−1 delay line taps are each downsampled by a factor of M. The Mdownsampled values are input signals to M allpass filters 401-0, 401-1,. . . , 401-(m−1). The filter output signals are summed in summer 402and scaled by 1/M scaler 403 to provide output samples y(m). Thetransfer function of the M-path polyphase IIR decimator 400 is given by:

${H(z)} = {\frac{1}{M}{\sum\limits_{k = 0}^{M}{z^{- k}{A_{k}\left( z^{M} \right)}}}}$Assuming that each of the M allpass filters 401-0, 401-1, . . . ,401-(m−1) has N cascaded real sections (i.e. N coefficients), thetransfer functions of the allpass filters 401-0, 401-1, . . . ,401-(m−1) each have the form:

${A_{k}(z)} = {\prod\limits_{j = 1}^{N}\;\frac{\beta_{k,j} + z^{- 1}}{1 + {\beta_{k,j}z^{- 1}}}}$Substituting the allpass filter transfer functions into the M-pathpolyphase IIR decimator 400's transfer function then gives:

${H(z)} = {\frac{1}{M}{\sum\limits_{k = 0}^{M}{z^{- k}{\prod\limits_{j = 1}^{N}\;\frac{\beta_{k,j} + z^{- M}}{1 + {\beta_{k,j}z^{- M}}}}}}}$Thus, the total number of filter coefficients in this example is M×N.

To illustrate some of the polyphase IIR decimators of the presentinvention, FIG. 5 shows a half-band, 2-path polyphase IIR decimatorstructure 500, configured in accordance with one embodiment of thepresent invention. As shown in FIG. 5, input samples x(n) are decimatedby 2 and applied as input samples to allpass filter 501. Concurrently,the input samples are delayed by one sample, decimated by 2, and appliedas input samples to allpass filter 502. Output signals of filters 501and 502 are summed in summer 503 and scaled by ½ at scaler 504 toprovide the output samples y(n).

Similarly, FIG. 6 shows third-band, 3-path polyphase IIR decimatorstructure 600, configured in accordance with one embodiment of thepresent invention. As shown in FIG. 6, input samples x(n) are decimatedby 3 and applied to allpass filter 601. The input samples are alsodelayed by one sample, decimated by 3, and applied to allpass filter602. Further, the input samples are delayed by two samples, decimated by3, and applied to allpass filter 603. The filter outputs of allpassfilters 601-603 are then added together by summer 604 and scaled by ⅓ toform output samples y(n).

FIG. 7 shows generalized 2-path polyphase IIR filter structure 700,configured in accordance with one embodiment of the present invention.As shown in FIG. 7, input samples x(n) are applied to allpass filter701, and filtered by P filter 703. The filtered samples are applied asinput samples to allpass filter 702. The filtered output samples ofallpass filters 701 and 702 are added together in summer 702 and scaledby ½in scaler 705 to provide output samples y(n).

To illustrate the transfer function of the allpass filters of thepresent invention, FIG. 8 shows 1-coefficient (“real section”) and a2-coefficient (“complex section”) IIR allpass filter structures 800 and850, respectively, configured in accordance with embodiments of thepresent invention. As shown in FIG. 8, for 1-coefficient filter 800, theinput samples are delayed in element 802 by one sample and added insummer 801 to a scaled difference (by β) between the input samples andsamples fed back from the output of IIR allpass filter 800. The samplesfed back from the output of IIR allpass filter 800 are delayed inelement 803 by one sample.

For 2-coefficient filter 850, the input samples are delayed two samples,and summed in summer 855 with the output samples of scaling element (byβ) 853 and the output samples of scaling element (by α) 854 to providethe filter output samples. The input samples to scaling element 853 arethe differences between the input samples of the 2-coefficient filter850 and the output samples of the filter, delayed by two samples. Theinput samples to scaling elements 854 are the differences between theoutput samples of the 2-coefficient filter, delayed one sample, and theinput samples of 2-coefficient filter 850, delayed one sample.

FIG. 9 shows 3-coefficient filter 900 which is formed by cascading threereal sections. In FIG. 9, delay elements (e.g., delay elements 903 and905) are shared between stages.

The texts by Lutovac and Milic, and Krukowski and Kale, discussed above,provide detailed descriptions of the theory and design of N-pathpolyphase IIR filters. These texts disclose algorithms for computing therequired allpass filter coefficients. In real world implementations,these filter coefficients are quantized to a finite number of bits.However, quantization by rounding or truncation results in significantfilter performance degradation (e.g., larger passband ripple and smallerstopband attenuation). The rounded or truncated coefficients aretypically far from optimal for the constrained number of bits.Algorithms for optimizing quantized filter coefficients include“bit-flipping” and “Downhill Simplex Method,” described in Chapter 3 ofthe Krukowski and Kale text, and “Differential Evolution” (DE) describedin the Storn text.

FIG. 10 illustrates quad-ratio (2, 3, 4, and 6) decimator multi-standardfilter 1000, configured in accordance with one embodiment of the presentinvention. As shown in FIG. 10, filter 1000 has an input sample rate ofF_(S) and an output sample rate which is selectable between F_(S)/2,F_(S)/3, F_(S)/4, and F_(S)/6, depending on the path selections throughthe filter structure.

For decimation by 2, the input samples bypass filter 1001 (with atransfer function labeled “H₁”) and decimation by 2 filter 1005, passthrough filter 1003 (with a transfer function labeled “H₂ ^(D2)”, withits associated decimation by 2) and filter 1004 (with a transferfunction labeled “H₃”). For decimation by 3, the input samples passthrough filter 1001 (labeled “H₁”), bypass decimation by 2 filter 1005,pass through the filter 1003 (labeled “H₂ ^(D3)”, with associateddecimation by 3), and pass through filter 1004 (labeled “H₃”).Similarly, for decimation by 4, the input samples pass through filter1001, decimation by 2 filter 1005, filter 1003 (with its associateddecimation by 2), and filter 1004. For decimate by 6, the input samplespass through filter 1001 and decimation by 2 filter 1004, filter 1002(with its associated decimation by 3), and filter 1004.

Table 1 shows examples of filters parameters for a multi-standard,multi-rate filter with component transfer functions H₁, H₂ ^(D2), H₂^(D3), H₃ ^(DVB-T), H₃ ^(ISDB-T) and H₃ ^(T-DMB), configured inaccordance with the present invention, for supporting DVB-T (also DVB-H,labeled “H₃ ^(DVB-T)”), ISDB-T (labeled “H₃ ^(ISDB-T)”), and T-DAB (alsoT-DMB, labeled “H₃ ^(T-DMB)”) applications. (The subscript of eachtransfer function indicates the transfer function's relative position tothe other transfer functions; for example, all the H₂ filters are inparallel relationship with each other, each receiving input samples fromthe output of the H₁ filter and providing samples to the H₃ filters). In

Table 1, frequency F_(PB) denotes the pass-band bandwidth, frequencyF_(SB) denotes the high-frequency edge of the stop-band, and F_(S)denotes the A/D sampling rate. Table 2 shows quantized filtercoefficients that satisfy the requirements shown in

Table 1, expressed as hexadecimal fractions and represent variouschoices of such coefficients for a multi-standard multi-rate filterconfigured in accordance with the present invention. In Table 2, A_(i)denotes the transfer function of the IIR filter in the ith path, andeach transfer function Ai may be implemented by multiple coefficients.

TABLE 1 Filter F_(PB)/F_(S) F_(SB)/F_(S) H₁ 0.104 0.385 H₂ ^(D2) 0.2080.271 H₂ ^(D3) 0.139 0.180 H₃ ^(DVB-T) 0.416 0.459 H₃ ^(ISDB-T) 0.3430.395 H₃ ^(T-DAB) 0.188 0.229

TABLE 2 A₀ Coefficients A₁ Coefficients ρ (β_(0i), α_(0i)) (β_(1i),α_(1i)) A₂ (β_(2i)) H₁ 44h/200h 127h/200h — H₂ ^(D2) — 8Ch/800h1EDh/800h 0h/800h 398h/800h 522h/800h 0h/800h 667h/800h 77Ah/800h0h/800h H₂ ^(D3) — 49h/800h F5h/800h 216h/800h 359h/800h 493h/800h5A6h/800h 683h/800h 746h/800h 7Beh/800h H₃ ^(DVB-T) 579h/800h 431h/800h,B5Eh/800h 538h/800h, C53h/800h — 65Eh/800h, D65h/800h 774h/800h,E69h/800h H₃ ^(ISDB-T) 333h/800h 209h/800h, 6EAh/800h 3C2h/800h,81Ah/800h — 593h/800h, 95Bh/800h 732h/800h, A79h/800h H₃ ^(T-DAB)−10Eh/800h 131h/800h, −264h/800h 372h/800h, −2FAh/800h 595h/800h,−388h/800h 73Eh/800h, −3F6h/800h

FIG. 11 illustrates quad-ratio (2, 3, 4, and 6) multi-standard decimatorstructure 1100, configured in accordance with yet a further embodimentthe present invention. FIG. 11 represents replacing each of the H₁, H₂,and H₃ filter blocks in FIG. 10 with the appropriate structure from FIG.4, FIG. 5 and FIG. 6, with all adder outputs truncated to 16-bits, andall paths scaled to ±1 except for the paths labeled (2), which arescaled to ±2, and the paths labeled (3), which are scaled to ±4.

FIG. 12 shows an example of the allpass filter sections 1104 and 1105 ofH₁ filter 1101 of FIG. 11, configured in accordance with the presentinvention. As discussed above, the adder outputs are truncated to16-bits and the paths are scaled to ±1 except for the paths labeled (2),which are scaled to ±2.

FIG. 13 shows an example of the allpass filter sections 1106, 1107 and1108 of the H₂ filter 1102 (i.e., H₂ ^(D2) and H₂ ^(D3) filters),configured in accordance with the present invention. As discussed above,the adder outputs are truncated to 16-bits, and the paths are scaled to±1, except for the inputs and outputs of each of the scaling elementsβ_(nm) (n running from 0 to 2 and in running from 1 to 3), which arescaled to ±2.

FIG. 14 shows an example of the allpass filter sections 1109, 1110 and1111 of the H₃ filter 1102 (i.e., H₃ ^(DVB-T), H₃ ^(ISDB-T), and H₃^(T-DAB) filters), configured in accordance with the present invention.As discussed above, the adder outputs are truncated to 16-bits and mostpaths are scaled to ±2 (as shown by the associated legends), except forthose scaled to ±1 or to ±4, as indicated.

Returning now to the zero-IF/very-low IF (ZIF/VLIF) radio receiverfront-end 100 shown in FIG. 1, receiver front-end 100 processesbroadcast digital multimedia signals in VHF and UHF bands. In thebroadcasting community, these frequencies are often referred to as BandI, Band II, Band III, Band IV, Band V, and L-Band.

Receiver front-end 100 is designed for the DVB-T/H, ISDB-T, and T-DABbroadcast digital multimedia standards, each of which uses an OFDMmodulation. The DVB-T/H channels are 5, 6, 7, and 8 MHz. The ISDB-Tchannels are 6, 7, and 8 MHz. The T-DAB channels are approximately 1.7MHz. By supporting T-DAB, the receiver front-end also supports the T-DMBstandard.

Receiver front-end 100 converts RF signals into quantized digitalsamples with minimal degradation. As shown in FIG. 1, the RF signal ismixed with in-phase and quadrature-phase local oscillator signals fromVCO 105 where the difference between the RF frequency and the VCOfrequency is either zero (ZIF) or very small (VLIF). The output signalsof mixers 103 and 1014 are each lowpass filtered by LPFs 107 and 108 toeliminate the higher frequency sum components, leaving the ZIF or VLIFcomponents.

To simplify filter implementation, thereby reducing die size (cost) andpower dissipation, and minimizing signal distortion, the lowpassbandwidth (B_(LP)) for LPF 107 and 108 may be selected to besignificantly larger than required by the signal bandwidth. For example,in one embodiment, the minimum lowpass bandwidth is one-half of themaximum channel bandwidth (e.g., 4-MHz for an 8-MHz channel). Onesuitable lowpass filter has a 60-dB bandwidth of 11.5 MHz. The filteredsignals are capacitively coupled to A/D converters 109 and 110 to removeany DC offset. A/D converters 109 and 110 sample and quantize the signalat sampling rate F_(S). The sampling rate should be sufficiently high toprevent aliasing, and be an integer multiple M of the OFDM signals' FFTsampling rates. Approximate sampling rates and integer multiples areshown in Table 3 for an 11.5 MHz B_(LP).

TABLE 3 Channel Bandwidth Standard (MHz) M F_(S) (Msps) DVB-T 8 2 18.3 72 16.0 6 3 20.5 5 3 17.1 ISDB-T 8 2 21.7 7 2 18.9 6 2 16.3 T-DAB 1.707 612.3

After the signals are sampled and quantized at A/D converters 109 and110, the samples are filtered and decimated by decimators 101 and 102 toobtain samples at the OFDM FFT sampling rate with minimal distortion.Decimators 101 and 102 each have a filter transfer function H(z) anddecimate by a factor of M.

Quad-ratio (2, 3, 4, and 6), multi-standard (DVB-T/H, ISDB-T, and T-DAB)decimator (QRMSD) 1100 of FIG. 11 is a suitable filter for implementingeach of decimator 101 and 102 of FIG. 1. Similarly, QRMSD 1100 is also asuitable filter for implementing QRMSD filters 301 and 302 of FIG. 3.

As discussed above, decimator 1100 is formed by cascading three filters1101, 1102 and 1103, having transfer functions H₁, H₂, and H₃,respectively. In some configurations, only one or two of the threefilters are required.

As shown above, transfer function H₁ can be configured as a half-band,decimate by 2 filter (H₁ ^(D2)), or as a low-pass filter (LPF) H₁, byswitching in, or out, additional delay elements without changing thecoefficients. Transfer function H₂ can be configured as either ahalf-band, decimate by 2 filter (H₂ ^(D2)), or as a third-band, decimateby 3, filter (H₂ ^(D3)), by changing the coefficients. Transfer functionH₃ can be configured as any one of a DVB-T LPF, H₃ ^(DVB-T), an ISDB-TLPF, H₃ ^(ISDB-T), or a T-DAB LPF, H₃ ^(T-DAB), by changing thecoefficients.

The H₁ filter may be designed to meet DVB-T's decimate-by-4 requirement(i.e., M=4), which are more stringent then those for the ISDB-T andT-DAB standards. Consequently, the same H₁ filter may be used for allthree standards. Further, an H_(i) filter designed for M=4 can also beused for M=6 and M=3, so that the same H₁ filter may be used for allstandards requiring decimations by 3, 4, and 6.

The H₂ filter may be designed to meet the DVB-T requirements, which areagain more stringent then those for the ISDB-T and T-DAB standards. Inthat manner, the same H₂ filter may be used for all three standards. TheH₂ ^(D2) filter is substantially identical for M=4 and M=2, and the H₂^(D3) filter is substantially identical for M=6 and M=3. Thus, only oneH₂ ^(D2) and one H₂ ^(D3) filter are required for all three standardsand all decimation ratios 2, 3, 4, and 6.

The H3 stop-band rejection filters are substantially identical for alldecimation ratios.

As mentioned above, the filter requirements are shown in

Table 1.

As shown in Table 2, single sets of H₁, H₂ ^(D2), and H₂ ^(D3) filtercoefficients, and three sets of H₃ filter coefficients are required tosupport all three standards and all four decimation ratios.

Thus, a multi-standard, multi-ratio decimator has been described.

The detailed description above is provided to illustrate specificembodiments of the present invention and is not intended to be limiting.Many modifications and variations within the scope of the presentinvention are possible. The present invention is set forth in thefollowing claims.

What is claimed is:
 1. A method for decimating a digital signal by afactor of M and matching it to a desired channel bandwidth, comprising:a. Applying input samples of said digital signal to an M-1 stage tappeddelay line; b. Downsampling said input samples and outputs of saidtapped delay line stages by a factor of M; c. Applying said Mdownsampled values to M allpass IIR filters where the phase of said Mallpass IIR filters add constructively at frequencies below a passbandfrequency, and add destructively at frequencies above a stopbandfrequency, and where: i. Said passband frequency is less than the inputsample rate divided by 2 times M; ii. Said stopband frequency is greaterthan the input sample rate divided by 2 times M; d. Summing the outputsof said M allpass IIR filters; e. Scaling said sum by a factor of 1/M;and f. Applying said scaled sum to a digital channel filter.
 2. Themethod of claim 1 where said allpass IIR filters are realized as1-coefficient structures.
 3. The method of claim 1 where said allpassIIR filters are realized as 2-coefficient structures.
 4. The method ofclaim 1 where said allpass IIR filters are realized as a cascade of1-coefficient structures.
 5. The method of claim 1 where said allpassIIR filters are designed using differential evolution.
 6. The method ofclaim 1 where said digital channel filter is a generalized N-pathpolyphase IIR filter.
 7. A method for selectively decimating a digitalsignal by a factor equal to the product of any number of the positiveintegers M₁, M₂, . . . , and M_(n), and matching it to a desired channelbandwidth comprising the steps of: a. Setting a buffer equal to saiddigital signal to be decimated; b. Setting k equal to 1; c. If M_(k) isin said product, inputting said buffer to the input of an M_(k)-pathdecimate by M_(k) method and placing output in said buffer; d.Incrementing said k; e. If k less than or equal to n, going to step c;f. Applying said buffer to a digital channel filter.
 8. The method ofclaim 7 where said M_(k)-path decimate by M_(k) methods each comprisethe following steps: a. Applying said buffer input samples to a M_(k)-1stage tapped delay line; b. Downsampling said input sample, and outputsof said tapped delay line stages, by a factor of M_(k); c. Applying saidM_(k) downsampled values to M_(k) allpass IIR filters each of whosephase responses add constructively at frequencies below a passbandfrequency, and add destructively at frequencies above a stopbandfrequency where: i. Said passband frequency is less than the inputsample rate divided by 2 times M_(k); ii. Said stopband frequency isgreater than the input sample rate divided by 2 times M_(k); d. Summingthe outputs of said M_(k) allpass filters; e. Scaling said sum by afactor of 1/M_(k).
 9. The method of claim 8 where said allpass IIRfilters are realized as 1-coefficient structures.
 10. The method ofclaim 8 where said allpass IIR filters are realized as 2-coefficientstructures.
 11. The method of claim 8 where said allpass IIR filters arerealized as a cascade of 1-coefficient structures.
 12. The method ofclaim 8 where said allpass IIR filters are designed using differentialevolution.
 13. The method of claim 7 where said digital channel filteris a generalized N-path polyphase IIR filter.
 14. The method of claim 7where said decimation ratio is selected based on measured ACI.
 15. Themethod of claim 7 where said decimation ratio is decreased when measuredACI increases.
 16. The method of claim 7 where said decimation ratio isincreased when measured ACI decreases.
 17. A method for decimating adigital signal by a factor of M and matching it to a desired channelbandwidth, comprising: applying input samples of the digital signal toan M-1 stage tapped delay line; downsampling the input samples of thedigital signal and output signals of each of the M-1 tapped delay linestages by a factor of M; applying the M downsampled input samples andthe M downsampled output signals of each of the M-1 tapped delay linestages to M allpass IIR filters where the phase of the M allpass IIRfilters add constructively at frequencies below a passband frequency,and add destructively at frequencies above a stopband frequency; andwherein the passband frequency is less than the input sample ratedivided by 2 times M; and wherein the stopband frequency is greater thanthe input sample rate divided by 2 times M.
 18. The method of claim 17,further comprising summing the outputs of the M allpass IIR filters. 19.The method of claim 18, further comprising scaling the sum of theoutputs of the M allpass IIR filters by a factor of 1/M.
 20. The methodof claim 19, further comprising applying the scaled sum to a digitalchannel filter.